To facilitate the reception of the signals transmitted by the satellites crossing large distances and being transmitted with limited powers, they are rendered as insensitive as possible to narrowband glitches by increasing their bandwidth by means of a spread spectrum technique by modulating the signal with a pseudo-random code.
The information to be transmitted at the level of the satellites, once put into the form of a series of frequency-spread binary data, is transposed into an emission frequency span by modulation with an emission carrier.
On reception, the binary information contained in a satellite radio signal of a positioning system is extracted by two demodulations.
The BPSK (Binary Phase Shift Keying) modulation technique consists in modulating, as may be seen in FIG. 1, the carrier of the RF (radio-frequency) signal transmitted by the satellite by a pseudo-random sequence known in advance of +1 and of −1, called a spreading code.
Represented in FIG. 1, from top to bottom, are the temporal evolution charts for the various components of a BPSK modulation radionavigation signal, namely, its spreading code, its carrier and the signal thus modulated, and then the autocorrelation function of the signal thus modulated and finally the spectrum of the modulated signal, that is to say the frequency distribution of its spectral density.
On reception at the antenna level it is possible to write:SBPSK(t)=cos(2π·Fpc·te+φ0)·code(te)
With:
te=te(t): date at which the signal received at the current instant t was transmitted
φ0: phase shift of the signal received
Or equivalently:SBPSK(t)=cos(φpc)·code(φcode)
BPSK modulation produces a cardinal sine power spectrum D(f) and a triangular autocorrelation function R(τ). The autocorrelation function R(τ) indicates the amplitude resulting from correlating the spreading code with a replica of this code delayed by τ. This function makes it possible to forecast the behaviour of the receiver which performs a correlation of the signal received with local replicas shifted by the spreading code so as to find the energy of the signal and perform a pseudo-distance measurement.
Certain navigation systems use double carrier shift signals consisting of an RF carrier (termed the “central carrier”) modulated both by a square sub-carrier and a spreading code. This modulation is called BOC modulation. This modulation exhibits a spectrum with two main components and an autocorrelation function with multiple peaks.
These signals are called BOC (“Binary Offset Carrier”) signals.
Represented in FIG. 2, from top to bottom, are the temporal evolution charts for the various components of a radionavigation signal received by a receiver, without double carrier shift defects, namely: its spreading code, the rectangular sub-carrier, the carrier and then the carrier thus modulated, and the autocorrelation function, and finally the frequency distribution of the spectral density of the modulated signal.
On reception at the antenna level it is possible to write:SBOC(t)=cos(2π·Fpc·Te+φ0)·sign(sin(2π·Fsp·te))·code(te)
With:
sign(x)=1 if x≧0 and sign(x)=−1 if x<0
Or equivalently:SBOC(t)=cos(φpc)·sign(sin(φsp))·code(φcode)
If the sub-carrier is regarded as a sinusoidal signal we may write:SBOC(t)≅cos(φpc)·sin(φsp)·code(φcode)i.e.SBOC(t)≅½[sin(φpc+φsp)−sin(φpc−φsp)]·code(φcode)
Or equivalently:SBOC(t)≅½·sin(2π·FpD·te+φ0)·code(te)−½·sin(2π·FpG·te+φ0)·code(te)
The two main components of BPSK type are retrieved with a carrier at the frequency FpD shifted to the right and a carrier at the frequency FpG shifted to the left.
Hereinafter, the following notation and definitions will be appropriate. The central carrier frequency Fpc is, by definition, the frequency of the carrier modulated by the code and the square sub-carrier is the frequency which is situated in the middle of the two carrier frequencies of the two components right Sd and left Sg.
The sub-carrier frequency Fsp is by definition the frequency of the square sub-carrier, which is equal to the distance between the carrier frequency of the left FpG or right FpD component of the spectrum and the central carrier frequency Fpc.
The frequency of the code is by definition the inverse of the duration of a code slot Lchip.
Hereinafter, the following notation will be used
Fpc Central carrier frequencyFpc=(FpG+FpD)/2
φpc is the phase of the central carrier receivedφpc=2·π·Fpc·te 
Fsp Sub-carrier frequencyFsp=−(FpG−FpD)/2
φsp phase of the sub-carrier received (rad)φsp=2·π·Fsp·te 
λsp wavelength of the sub-carrierλspr=c/Fsp 
FpD Carrier frequency of the right component
FpG Carrier frequency of the left componentFpG=Fpc−Fsp FpD=Fpc+Fsp 
Fcode Frequency of the spreading code
φpc is the phase of the code receivedφcode=te 
λcode is the wavelength of the spreading codeλcode=c/FcodeLchip=1/Fcode
φcode is the phase of the spreading codeφcode=Fcode·t 
BOC modulation has a double aim:                to free the spectrum between the two components for other already existing signals,        to improve the precision of the measurements in the presence of thermal noise and multi-paths.        
The drawback of this modulation is that in order to correctly demodulate the signal, it is necessary to find the main peak of the autocorrelation function so as to have the maximum of energy and to provide coherent measurements between the satellites, otherwise biased pseudo-distance measurements are provided.
There exist methods of the BPSK type (termed “BPSK like” methods) in which, in a transition phase, the two components left and right of the spectrum of the double carrier shift signals dispatched by the satellites are demodulated in parallel as if each component were a conventional BPSK signal, with no local sub-carrier, each of these components having a carrier shifted to the left or to the right. The processing is performed in parallel for the signals originating from the various satellites.
The demodulation conventionally consists in correlating in parallel the components generated locally with the components received from the satellite considered. Combination of the local components forms a local signal comprising a local code and a local carrier.
The receiver seeks to slave in phase, by means of a code loop and a central carrier loop, the local codes and, respectively, the local carriers of the local components, with the codes and, respectively, the carriers of the two components received by searching for the maximum of the autocorrelation function (in the BPSK like mode, the autocorrelation function corresponds practically to the envelope of the autocorrelation function of the double carrier shift signal).
Once the code loop has converged on the maximum of the envelope of the autocorrelation function, the receiver undertakes a tracking phase. In the tracking phase, the main peak of the autocorrelation function of the BOC signal is followed in BOC mode or in reconstituted BOC mode by means of code and central carrier loops.
In the BOC mode the signal is not decomposed into two components, right and left.
In the tracking phase in reconstituted BOC, the two components left and right are demodulated, in BPSK mode, the demodulated left and right complex components are summed in a coherent manner and the central carrier and code loops are closed.
The measurements of pseudo-speed and pseudo-distances, are formulated on the basis, respectively, of the phase of the local carrier and of the phase of the local code during the tracking phase.
However, an ambiguity may persist while switching to the tracking phase. There is no certainty that the fact of positioning ourselves on the maximum of the envelope of the autocorrelation function in the transition phase, leads us, in the tracking phase, to follow the main peak of the autocorrelation function of the signal in reconstituted BOC or BOC mode.
When the signal possesses the ideal, perfectly symmetric form, as is the case in respect of FIG. 2, the autocorrelation function R(τ) always exhibits a predominant main peak at the centre and secondary peaks of lesser amplitude on either side. In this case, it is relatively easy, by comparing the amplitudes or by utilizing the envelope of this function, to pinpoint the main peak of this function with high confidence.
However, when the signal is deformed by the analogue pathways (non-ideal transfer functions on the antenna, the filters, the amplifiers and the frequency changing analogue multipliers) within the receiver or within the satellite, it is possible to obtain a non-symmetric autocorrelation function, or indeed in the worst case one which is anti-symmetric with two main peaks on either side of the centre, of opposite signs, as represented in FIG. 3a. In this case, it is difficult, or indeed impossible, to make the choice to switch to nominal tracking mode based on a criterion of maximum amplitude of the envelope of the autocorrelation function. Furthermore, energy is lost with respect to the symmetric case.
This phenomenon is due to an incoherence between the relative phase shift of the two components on one side and the mean group delay on the two components on the other. This incoherence is due to a non-constant group delay in the passband (or stated otherwise a phase delay which is non-linear in frequency). This defect is called “phase differential”. An example of this phenomenon has been represented in FIG. 3b. In this figure have been represented, from top to bottom, the graphs, as a function of frequency, of the group delay, of the phase delay and of the spectral distribution of the energy of the signal exhibiting these defects. The same phenomena occur when defects affect the signals transmitted by a satellite.
The presence of multi-paths can also bias the convergence of the code in BPSK mode and induce a false lock-on to a secondary peak in the tracking phase if the code error, that is to say the phase difference between the local code and the code of the signal received exceeds, on input to the tracking phase, half the distance between two consecutive peaks of the autocorrelation function of the BOC signal.
In the ideal case where the limited-band filters at transmission and at reception let through all the secondary components of the signal that lie between the 2 main right and left components of the spectrum without deforming them (ideal template), the form of the autocorrelation function in BPSK demodulation remains sufficiently triangular to allow, before the switch to tracking mode, convergence of the code to the main peak of the autocorrelation function of the BOC signal even in the presence of multi-paths.
However, the analogue filters which are non-ideal at transmission and at reception round the form of the autocorrelation function, thereby increasing the sensitivity to multi-paths during the convergence of the code in the transition phase, with a real risk of false lock-on to a secondary peak of the autocorrelation function in the tracking phase.
In order to limit the risks of false lock-on to a secondary peak of the autocorrelation function during the tracking phase, methods are known for resolving ambiguity (ie removing ambiguity), during the transition phase which precedes the tracking phase, which are suitably adapted for making the code converge, before tracking, onto the main peak of the autocorrelation function of the signal received.
The object of the method of resolving ambiguity is to generate a local code which is sufficiently close to the code received, before the switch to tracking mode, to ensure that the tracking is done on the main peak of the autocorrelation function of the BOC signal. It will be considered hereinafter that the code error is zero when we are at the top of the main peak of the autocorrelation function of the BOC signal.
In the article “Acquisition of the PRS BOC(15,2.5) Signal in Presence of Multipath” by Martin, N; Guichon, H; Revol, M; Hollreiser, M; Crisci; there is proposed a method for resolving ambiguity so as to avoid false lock-ons to a secondary peak, due to multi-paths.
We shall first of all describe a digital processing channel of a receiver, adapted for implementing a method of resolving ambiguity for a BOC signal transmitted by a satellite of index i. The signals transmitted by the satellite are, prior to processing by the digital processing channels, received and digitized by the analogue processing pathways. A digital processing channel is represented in the form of a block diagram represented in FIG. 4. A digital processing channel is able to process signals received 1i originating from one and the same satellite i.
The spectrum of a BOC signal comprises two spectral components Sdi and Sgi, as visible in FIG. 2. A BOC signal is regarded as a signal having two BPSK components with an identical code but two distinct carriers, a deterministic and known relation existing between the transmit phases, before the defects due to the analogue part of the receiver (and possibly of the satellite).
The digital processing channel of the satellite of index i exhibits a hardware correlation pathway 50 able to generate the local codes and the local carriers of the right and left components of the local signal and to correlate, thereafter, these local components with the right and left components of the signal received. The local signal itself comprises a local code and two local carriers right and left and a local central carrier.
The software channel 40 comprises a code loop DLL and a central carrier loop PLL which are able to identify phase deviations between the local code and the code of the signal received and respectively, the local central carrier and the central carrier received. The code and central carrier loops thereafter generate commands for controlling the hardware correlation pathway 50 and generating a new local signal.
The central carrier phase loop PLL makes it possible to ensure precise tracking of the dynamics of the phases of the signals (due to the motion of the antenna, to the drifting of the clock of the receiver and to the displacements of the satellites).
The hardware correlation pathway 50 comprises two left G and right D correlation channels, to carry out the demodulation in BPSK mode. The two left G and right D correlation channels are able to independently demodulate, respectively, the two spectral components right Sdi and left Sgi of a radionavigation signal originating from a satellite of index i. In FIG. 4, the thick lines represent complex signals and the thin lines real signals.
The left G and right D correlation channels comprise means for correlating the right Sdi and left Sgi spectral components of a signal received 1i with the right and left local components of a signal produced locally so as to produce complex outputs of the right ZAD, ZPD, ZRD and, respectively, left ZAG, ZPG, ZRG correlation pathways. These outputs are different for each digital processing channel of index i but for greater clarity, the index i is not indicated on these outputs.
The left and right correlation means each comprise a multiplier 2g, 2d whose output is linked to a set 3g, 3d of three correlation multipliers placed in parallel, whose outputs are linked to a correlation integrator 4g, 4d. 
The integrators 4g, 4d have the role of producing samples of demodulated, despread left ZAG, ZPG, ZRG (advance, punctual, delay) and right ZAD, ZPD, ZRD signal at low rate, for example, every 20 ms, at the software part 40 on the basis of the products at the output of the code demodulators formulated at high rate.
The term code demodulator designates the multiplier between a complex carrier-demodulated component received and a local code.
The term complex correlator designates the set formed by a code demodulator and an integrator 4g, 4d with periodic resetting to zero. The resulting complex product is the despread demodulated signal received. Here we have three complex correlators per correlation pathway.
The hardware pathway 50 also comprises a code generation circuit 24 able to generate and to provide right advanced CAD, punctual CPD, delayed CRD and left advanced CAG, punctual CPG, delayed CRG local codes to the right D and left G correlation channels, on the basis of code commands CC and of carrier commands CP.
The code generation circuit 24 comprises a code phase digital integrator NCOc 18 controlled by speed code commands CC. The speed code commands are previously amplified by means of a speed code commands amplifier 67. The central-carrier speed commands CP are added to the code-speed commands by means of an adder 29, before the speed code commands amplifier 67. The carrier loop is said to help the code loop so as to reduce the trailing of the code loop due to dynamics, thereby making it possible to reduce the code loop band and therefore the noise in the measurement of the phase of the code for the pseudo-distance measurement.
The code phase digital integrator NCOc, 18 is able to generate a local code phase φcodeL. It involves a digital integrator (without reset to zero) which produces the phase of the local code at high frequency (˜100 MHz) on the basis of speed and jump commands updated by the software at low frequency (˜50 Hz).
The code correlation circuit 24 comprises a local codes generator 19 controlled by the local code phase φcodeL and providing right CAD, CPD, CRD and left CAG, CPG, CRG local code replicas to the respective right D and left G correlation channels.
The hardware pathway 50 furthermore comprises a carrier correlation circuit 25 able to generate and to provide and right Pld and left Plg local complex carriers, to the right D and left G correlation channels on the basis of the central carrier speed commands.
The commands, the replicas of the local codes and the local carriers are produced for each digital processing channel of index i but, for greater clarity, the index i of the satellite for these data is not indicated in the text of the patent application.
The carrier correlation circuit 25 comprises a local central carrier phase digital integrator NCOp 9 controlled by central carrier speed commands. The carrier commands CP are previously amplified by means of a second amplifier 8.
The local central carrier phase digital integrator NCOp 9 generates the local central carrier phase φpcL. The local sub-carrier phase φspL, obtained after amplification of the local code phase φcodeL by means of a sub-carrier amplifier 22, is added to and simultaneously subtracted from the local central carrier phase φpcL to produce respectively two local carrier phases right φpDL and left φpGL. For this purpose, the hardware pathway 50 comprises a right adder 10 summing the local central carrier phase φpcL and the local sub-carrier phase φspL so as to obtain a right local carrier phase and a left subtracter 12 subtracting the local sub-carrier phase φspL from the local central carrier phase φpcL so as to obtain a left local carrier phase φpGL.
The hardware pathway 50 furthermore comprises right 11 and left 14 carrier generators, generating the right Plg and left Pld complex local carriers on the basis of the right φpDL and left φpGL local carrier phases. The local carriers thus produced are dispatched to the left and right correlation channels.
The code loop DLL comprises a code discriminator DSR, 26, making, on the basis of signals arising from the complex outputs of the left and right correlation pathways which are transmitted to it every 20 ms, estimations of the code error ε0i. The code discriminator provides the instantaneous estimations ε0i of the code error to a code corrector CRC 15. The code corrector CRC, 15, uses these estimations to generate speed code commands CC every 20 ms. The code error εi for a satellite i represents the difference between the phase of the local code and that of the code of the signal received from the satellite of index i.
The central carrier phase loop PLL comprises a carrier discriminator DSP, 27 making, on the basis of signals arising from the complex punctual outputs of the right ZSPD and left ZSPG correlation pathways, an estimation of the central carrier phase error θ0i and providing the latter to a central carrier corrector CRP, 7, generating central-carrier commands CP in the form of carrier speed.
The central carrier phase error θ0i represents the difference between the phase of the central carrier of the signal received and the phase of the central carrier of the local signal.
The signals, arising from the complex outputs of the right ZSAD, ZSPD, ZSRD and left ZSAG, ZSPG, ZSRG correlation pathways, used by the carrier and code discriminators are here the complex outputs of the right ZAD, ZPD, ZRD and left ZAG, ZPG, ZRG correlation pathways.
The method of locating a mobile craft by radionavigation, described in the article cited previously, comprises a series of steps. First of all an energy-based search step is undertaken. The search phase consists in detecting the signal transmitted by the satellite of index i by searching for a correlation peak between the local signal and the signal received. In this search phase, the two components right and left of the signal received are demodulated in parallel by means of the hardware pathways 50 such as described previously by scanning, in open loop, several hypotheses about the phase of the code and about the frequency variation of the central carrier called the Doppler. The energy at the output of the correlators is measured. When an energy greater than a predetermined threshold is detected, then the signal is present.
The receiver thereafter undertakes a transition phase the object of which is to make the code converge towards the maximum of the envelope of the autocorrelation function in BPSK mode (namely the maximum of the envelope of the autocorrelation function of the BOC signal). For this purpose, the right and left components of the signal received are demodulated in parallel, in BPSK mode, by means of the hardware correlation pathway 50 and frequency and code loops, not represented, are closed. The frequency loop slaves the frequency of the local central carrier to the central carrier frequency of the signal received.
A phase of ambiguity resolution is carried out thereafter so as to make the code converge, in mode BPSK, onto the main peak of the BOC autocorrelation function of the signal received by means of the digital processing channel represented in FIG. 4.
The method of resolving ambiguity proposed in the aforementioned article is carried out in parallel on the signals arising from the various satellites in the digital processing channels such as previously described. For the signals originating from a satellite of index i, the following processing is carried out until the code loop converges:
The right and left components of the signal received are demodulated by a conventional BPSK demodulation method by means of the right and left channels. A code loop and a central carrier phase loop are closed so as to generate central-carrier commands and code commands after having made estimations of the code errors ε0i and central carrier phase errors θ0i.
New local codes and new local central carriers for the two right and left local components are generated by means of the code generation 24 and central carrier 25 circuits on the basis of the central-carrier and code commands and BPSK mode demodulation is carried out again as long as the code loop has not converged.
The proposed method of resolving ambiguity comprises two successive steps. In a first step, the code loop is made to converge with a correlation circuit of tight correlator type until the standard deviation of the code error is less than a first code convergence threshold. On completion of this step the risk of false lock-on to a secondary peak still exists since the code error may despite everything be greater than half the distance between two peaks of the autocorrelation function of the BOC signal, because of the possible presence of multi-paths.
In a second step, the code loop is again made to converge, with a correlation circuit of Double Delta type. The code can then converge until the standard deviation of the code error is less than a second code convergence threshold. On completion of this step, the error in the code is assumed to be less than half the distance between two peaks, which is typically 5 m for a BOC signal generated by the Galileo system. The risk of false lock-on to a secondary peak, during the subsequent tracking phase, is markedly reduced.
On completion of the ambiguity resolution phase, a BOC mode tracking phase is entered, with a single correlation pathway and a single square local sub-carrier.
The satellite location system calculates pseudo-distances on the basis of code phase values arising from the tracking step.
The method of resolving ambiguity described exhibits the advantage of ensuring, with good robustness to multi-paths, that the nominal tracking is done on the main peak of the autocorrelation function.
However, the estimations ε0i of the code errors on which the code corrector is based are noisy and biased by multi-paths, thereby inducing a non-negligible risk of false lock-on to a secondary peak. The consequence of this is to not ensure the integrity of the measurements of pseudo-distances and pseudo-speeds carried out by a location method based on a method of resolving ambiguity according to the prior art.
The aim of the invention is to remedy the aforementioned drawbacks.